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The Audio Signal Path; Minimising Power Supply Interaction

June 25, 2011


The comment “the sound of the rectifier” is always an alert to me. I assert that if the rectifier is audible, then there is signal current flowing in the least linear part of the circuit, the power supply and in particular, the rectifier! I have been on an explorative and experimental path looking to minimise the degree to which signal currents flow in the power supply.  The first place to look is the output stage since the current requirements of that stage are the most demanding (usually). In considering the output stage, the first issue that comes up is the class of operation. In this respect, Class A has an advantage and so in this discussion we are limiting ourselves to Class A operation. However, simply limiting our designs to Class A is not enough. The problem is that the current demands of even a Class A stage will change according to the signal level. For example, consider the data sheet case of a 45 triode operating single-ended in Class A at 250 V with a bias current of 34 mA, delivering 1.6 W into 3.9 kΩ:

Placing a 3.9 kΩ load line on the curves passing through 250 V at 34 mA, we can note that the plate currents at the ends of the line as defined by a grid swing of ±49.5 V are at 67.5 mA and 6 mA, the mean of which is 36.75 mA.

We can see that this is elevated 2.75 mA compared to the quiescent current of 34 mA; this is due to the departure of the tube from perfect linearity and is sometimes referred to as rectification; that is to say, the current in one direction is greater than the current in the opposite direction; a diode simply takes this difference in “back-to-front” current to the limit.  When reproducing music we rarely if ever, encounter steady state conditions (except for silence) and so this example is somewhat unrealistic however, it does serve to illustrate that even in Class A, the signal will place demands on the power supply. So, given that no amplifying device is perfectly linear, what can we do to minimise audible sonic interaction of the signal with the power supply?

The traditional approach is to use a combination of chokes and capacitors acting as an energy reservoir to satisfy the current demands resulting from the signal acting upon a less than perfectly linear output tube. That experimenters and observers often note “the sound of the rectifier” indicates that such traditional measures are not adequate in the quest for sonic fidelity. This article shows how the signal may be isolated from the power supply using a simple and practical technique.


First, let us review the signal current loops in a cathode biased, single ended output stage, referring to figure 1:

The first loop is the input loop (Dashed);

What is not commonly stated is that an amplifying device responds to a voltage difference in this case, the difference between the grid voltage and the cathode voltage. If the grid voltage falls relative to the cathode voltage, the current passed by the tube falls and vice-versa. With a common cathode stage, the cathode is held at signal ground potential by the cathode bypass capacitor while the grid ‘sees’ the input signal. The tube grid input impedance is not infinite, in fact RDH4 has this to say with regard to grid input impedance: “When a valve is used at low audio frequencies, it is sometimes assumed that the grid input impedance is infinite. In most cases however, this assumption leads to serious error, and careful attention is desirable to both its static and dynamic impedances.” For the purpose of our discussion, the serious error referred to is that the input signal has no current associated with it such that the input loop is usually not considered carefully; the point being that if a cathode bypass capacitor is used this current flows from the source to the grid, to the cathode and via the cathode bypass capacitor back to the source and vice versa, alternating with the signal: the cathode bypass capacitor is in the input signal path.

The second case is the output loop (Chain Dashed);

Similarly to the input case, the output voltage develops across two terminals of the amplifying device, in this case the plate and the cathode. In the case of a transformer coupled load, the voltage acting across the load reflected to the primary causes current to flow from the plate to the output transformer primary, to the PSU bypass capacitor to ground and via the cathode bypass capacitor to the cathode and vice versa, alternating with the signal current. Note that there are two capacitors in series with the output signal current! (We will address this issue later in this article.)

The third case is the power supply (PSU) loop (Dotted);

In the case of a perfect Class A amplifier, there would be no power supply signal loop, the power supply would merely be feeding the circuit with energy to balance the energy dissipated in the load. Furthermore, if the PSU bypass capacitor were perfect (zero impedance at all frequencies) then the power supply would not be in the signal path however, there is no such thing as a perfect Class A amplifier or a perfect capacitor and so the power supply is in the signal path. To make matters worse, the energy supplied by both the power supply and PSU bypass capacitor has to pass via the cathode bypass capacitor.

We have identified two signal path issues that audibly impact the sonic results of the circuit: One; Signal current flowing in the power supply and Two; The cathode bypass capacitor that appears in all three loops; the input loop, the output loop and the power supply loop.

Taking the power supply first; the traditional approach is to simply increase the energy storage thereby increasing the period of time averaging. The obvious issue here is that the equivalent series resistance (ESR) of a capacitor is not flat with respect to frequency and even more, the impedance of the power supply as a whole is far from flat with respect to frequency. This means that the impedance of the power supply signal path is not flat and so the power delivered into the load is not flat. Note, even if the measured performance appears flat, our ears are quite capable of distinguishing micro-effects that limit the potential musical beauty of an amplifier and that is, I assert, the point of this hobby so long as we are having fun. Worse still, many L-C or C-L-C power supplies exhibit a severe resonance (impedance peak) in the region 5 to 15 Hz. In fact, all power supplies are to some degree reactive (i.e. they resonate) and this has a disastrous impact on the sonic results that some people may actually like, however, it is not conducive to fidelity.

When I started designing tube amplifiers (around 12 years ago), I was aware of the power supply resonance issue and my approach was to employ tube based voltage-regulated power supplies and this is quite effective. The advantage of a regulator is that there is a substantial voltage drop across it that corresponds to available stored energy, much lessening the degree to which signal current flows in the rectifier. I then started using a simple mosfet series regulator with a reference voltage on the gate. (I also put a long RC time-constant on the gate to obtain slow turn-on.) This is highly effective, affording notably flat power source impedance in the 6 Ω region. What I noticed is that the sound improved over that of relatively sophisticated error compensating tube power supplies. However I felt I could do better. The problem is that the power supply signal loop is now a composite of the regulator and the bypass capacitor. The tube regulators I built have source impedances in the 1 to 4 Ω region. This means that up to 1000 Hz or so*, I was listening to the regulator while above 1000 Hz, the PSU bypass capacitor dominated the sonic results. “Listening to the regulator” is a common and valid objection to voltage-regulated power supplies. The series mosfet may help by eliminating the error amplifier. Even so, we are still left with the problem of the rectification effect of the amplifier tube causing current demands that are not adequately managed by the power supply in sonic terms.

*For a PSU source impedance of 4 Ω, the bypass capacitor would have to be 4000 µF to be effective down to 10 Hz. Apart from not being practical at tube power supply voltage levels, such a large capacitor (or bank of capacitors) is likely to sound gruesome.

Many of you may find the use of solid-state devices bothersome for various reasons. I assert that there are a few ways in which solid-state devices free us to explore topologies that, while not necessarily new, are not practical when using tubes only. In fact, used with due attention to detail, there is a particular solid-state device that permits fuller realisation of the inherent sonic beauty of vacuum tubes; The solution to the power supply signal interaction problem that I am proposing is to harness depletion mode mosfet technology to create truly effective isolation of the signal from the power supply.

I did a simple investigation of the performance of the two-terminal depletion mode mosfet cascode CCR, see figure 2. Note, I am using the initials CCR, viz; constant current regulator, since this device is equally effective as a source or a sink.

What I found was that using the Supertex DN2540 for M1 and M2, the measured DC current variation was less than 0.5 mA over a range of impressed DC voltage from 50 V to 200 V and the variation was immeasurable over a 100 V range. This is extraordinary performance for such a simple two-terminal circuit.

Some people report that for plate or cathode “load” applications, the IXYS 01N100D is superior sonically, I have not attempted such a comparison; I did repeat the DC current test using the 01N100D for M1 and M2 and the performance was not acceptable. I then replaced M1 with the DN2540 and found the resulting DC performance to be acceptable but not quite as solid as that obtainable when using the DN 2540 for both M1 and M2.

The AC test set-up used one section of a 6AS7 as a cathode follower, with the CCR (using the DN2540 for both M1 and M2) in the cathode circuit, see figure 3.

A test load of 500 Ω was inserted between the CCR and the negative rail. I drove the 6AS7 such that 20 VRMS (VIN) was measured at the cathode and then measured the voltage (VOUT) at the load. I ensured that Vin was constant for each test frequency. The AC impedance of the CCR is very sensitive to the value of gate stopper resistor; I chose to implement the design using the same value chosen by Walt Jung, 100 Ω1. (It is worth noting that I have not experienced oscillation of the CCR devices I have built using 100 Ω gate stopper resistors.) My results are very different than those of Walt Jung, probably due to a different test method and much higher current combined with a large input voltage; however, my set-up represented my application; at a test current of 65 mA, the cascode CCR exhibited an impedance greater than 1.5 MΩ over a range of 10 Hz to 40 kHz.

This circuit is quite remarkable and opens the door to new, extreme-performance tube audio implementations that were not possible before the advent of depletion mode power mosfets. My current (ahem) preferred choice of devices is the DN2540 for M1 and the 01N100D for M2. Note that the 01N100D is rated for 1000 V, whereas the DN2540 is rated for 400 V; Since the VDS of M1 is limited by the VGS of M2, the voltage withstanding capacity of the cascode CCR is determined by the voltage rating of M2.



Given that a CCR has an impedance of 1.5 MΩ or greater over and beyond the audio spectrum, placing a CCR in series with the plate supply will result in almost perfectly constant current; that is, the power supply will not be modulated by the signal and will simply feed energy to the circuit to balance the energy dissipated in the load; this is the ideal we are striving for. To justify this statement, let us take a look at some numbers:

We will continue with our 45 triode operating single-ended in Class A at 250 V with a bias current of 34 mA having a plate resistance 1.75 kΩ and a reflected load of 3.9 kΩ.

We first must establish the resistance of the stage at the lowest frequency of interest, say 5 Hz. Let us also say we have an output transformer having primary inductance of 40 H.

The impedance at 5 Hz is given by 2 x π x f x L = 2 x π x 5 x 40 = 1256 Ω.

In parallel with this we have the reflected load, in our 45 case, 3.9 kΩ resulting in a composite load of 1.256 kΩ in parallel with 3.9 kΩ = 0.95 kΩ.

In series with this we have the plate resistance, again for our 45 case; 1.75 kΩ and the cathode bypass capacitor, let’s say for the purpose of discussion 2 Ω, that is negligible and so we will ignore it. The resulting stage resistance is 0.95 kΩ + 1.75 kΩ = 2.7 kΩ.

Let the plate signal current be IP. If there were no PSU bypass capacitor, the voltage due to the signal at the PSU end of the output transformer would be IP x 2.7 kΩ (neglecting the Thevenin effect of the CCR and PSU impedance since 1.5 MΩ >>> 2.7 kΩ).

The PSU source impedance and the CCR impedance form a potential divider, the top leg being the CCR impedance and the bottom leg being the assumed PS source impedance of 2 Ω.

For a potential divider, VOUT = VIN x RBOTTOM ÷ (RTOP + RBOTTOM) and so we have the voltage at the CCR / PSU node is;

VCCR/PSU = IP x 2.7e3 x 2 ÷ (1.5e6 + 2) which simplifies to IP x 3.6e-3.

To calculate the power supply rejection ratio, we simply divide the voltage at the CCR / PSU node by the voltage at the PSU end of the output transformer giving:

PSRR = IP x 3.6e-3 ÷ IP x 2.7e3 = 1.333e-6 or -117.5 dB, an excellent result!

Note; If you are following closely, you will have noticed that the stage resistance cancels and that the PSRR may also be calculated by dividing the PSU source impedance by the CCR impedance. If we are pedantic, we will include the PSU source resistance in the denominator but 1.5 MΩ >>>> 2 Ω so we will neglect it. This gives the power supply rejection ratio as 2 ÷ 1.5e6 = 1.333e-6 or -117.5 dB, the same figure we had calculated from first principles and so it can be seen that the power supply rejection ratio is dominated by the impedance of the CCR. This analysis illustrates and confirms the effectiveness of CCR as a powerful tool in our quest to isolate the signal from the PSU.

In fact, 117 dB is close to or beyond the dynamic range of our hearing so it is possible that we have achieved our ideal, any signal current modulation of the power supply will now be inaudible and the power supply is now doing the only thing that we need it to do, that is supplying the energy dissipated by heating of the circuit elements and in the load. Note, even if the power supply impedance rises to 100 Ω (quite possible if resonance is present), the power supply rejection ratio is dominated by the CCR and will still be -83 dB.

Now that we have effectively removed the power supply loop from the signal output loop, the output signal loop completion is simplified to the PSU bypass capacitor; what value should this capacitor be?

The signal output or load loop is comprised of the stage resistance (already calculated as 2.7 kΩ), the PSU bypass capacitor to ground and then via the cathode bypass capacitor to the cathode. (It should be noted that the PSU capacitor and the cathode bypass capacitor appear in series the consequences of which we examine later.)

The minimum value of the bypass capacitor is given by C = reciprocal (2 x π x f x R).

Again, our lowest frequency of interest of is 5Hz and so we have

Reciprocal (2 x π x 5 x 2.7e3) = 11.78 say 12 µF.

This value is available in high quality film devices; we no longer need to horribly compromise our lovingly created signal path with an electrolytic capacitor. I usually parallel this with an excellent coupling capacitor such as an Auricap of not more than 1/100 value; in this case 0.1 µF would be a good choice.

Now, let us take a look at the cathode bypass capacitor:

To bias the 45 according to our example, the cathode bias resistor will be 50 V / 45 mA = 1.11 kΩ. Thevenin tells us that this resistance appears in parallel with the cathode resistance, given by 1/Gm. In our case, the data sheet gives Gm as 2.175 mA/V; taking the reciprocal, we get the cathode resistance as 0.46 kΩ. 0.46 kΩ in parallel with 1.11 kΩ gives 0.325 kΩ. Again, taking the lowest frequency of interest to be 5Hz, the value of the cathode bypass capacitor is reciprocal (2 x π x 5 x 325) = 100 µF.

As we have already noted, this capacitor appears in series with the PSU bypass capacitor thus reducing the PSU bypass capacitance to reciprocal (1/12 + 1/100) = 10.7 µF. Not too bad and increasing the PSU bypass capacitor to 13.6 µF (say 15) will serve our design intentions. Please note however, that the cathode bypass capacitor not only appears in the signal input loop but also in the signal output loop in series with the PSU bypass capacitor (again, refer to 1). No capacitor has flat ESR with frequency and the larger the capacitor, the worse the ESR behaviour and connecting two large capacitors in series makes matters worse. This is why I am a staunch advocate of grid bias* (or, in the case of voltage gain and drive stages, LED bias). The sonic impact of cathode bypass capacitors is far from theoretical; it is very audible.

(*Note, I distinguish grid bias from fixed bias: Grid bias is simply applying a bias voltage to the grid. Combining grid bias with fixing the plate voltage results in fixed bias. As I describe later, I set the grid voltage, set the current using a CCR and let the plate voltage float.)

The PSU bypass capacitor will increase the power supply rejection ratio since the instantaneous signal current will be sourced by the bypass capacitor. The effect of the capacitor is to drastically reduce the voltage due to the signal at the PSU end of the output transformer; this will now be given by the plate signal current multiplied by 2.7 kΩ in parallel with the ESR of the capacitor, say 2 Ω; since 2.7 kΩ  >>> 2, Ω let us say 2 Ω. To be pedantic, we should also take account of the CCR resistance and the power supply source impedance that is, 1.5 MΩ + 2 Ω. Again, since 1.5 MΩ  >>>> 2 Ω, the effect is negligible and so we will use 2 Ω giving the voltage at the PSU end of the output transformer as IP x 2. This can be compared with the result previously obtained of IP x 2.7 kΩ.

As before, the PSU source impedance and the CCR impedance form a potential divider, the top leg being the CCR and the bottom leg being the assumed PS source impedance of 2 Ω.

Again using the potential divider equation, the voltage at the CCR / PSU node will now be (IP x 2) x 2 ÷ (1.5e6 + 2) which simplifies to IP x 2.667-6.

Again, to obtain the power supply rejection ratio we have:

PSRR = IP x 2.667-6 ÷ IP x 2 = 5.334-7 or -125.5dB.


Figure 5A shows a parallel feed CCR topology that is a Western Electric concept I came across in Lynn Olson’s presentation “Loop Distortion”2. In this case the CCR is acting as a gyrator while the output transformer is parallel fed by a capacitor that closes the output loop directly to the cathode. The classical version of this circuit uses a choke in place of the CCR. The choke provides a large degree of isolation from the PSU; this may be one reason why parallel feed amplifiers have a strong following. However, a choke may not provide the same level of signal isolation and noise attenuation from the PSU that a CCR can, both at the LF end due to limited reactance and at the HF end due to shunt capacitance. It is worth noting that when this WE topology was conceived, the CCR would have been tube based that, in requiring a much elevated B+ voltage together with added parts cost, would have had limited commercial viability. Now, with the advent of depletion-mode mosfets, this topology is not only attractive, but practical too. My implementation uses direct feed, see figure 5B; I am considering making it switchable on the fly between parallel feed and direct feed for the sake of comparison and fun! Practically speaking the switching is a simple matter of moving the CCR connection between ends of the output transformer primary perhaps using wetted relays.

You may observe that both the above topologies look like the Ultrapath topology trademarked by Jack Elliano3, the difference is that Ultrapath has no specific provision to isolate the signal from the power supply; it relies on a conventional PSU bypass capacitor. It is also worth noting that a cathode feedback capacitor is not shown in either case and may not be necessary; it is possible however, that audible levels of noise may be injected at the cathode. Fixing this gets us back to having a big capacitor in the input signal loop and I would advocate the use of grid bias with a coupling transformer to avoid this pitfall.


Figure 6 shows the topology of my recent direct-feed single ended 50-300B project. It is extremely simple, particularly with respect to the signal paths since the power supply has been removed from the signal path by the CCRs. CCR power supply technology may be combined with grid bias for current gain / output stages or LED bias for voltage gain / drives stages to address both the signal path compromises identified at the beginning of this article; accordingly this design uses LED bias at the drive stage and grid bias at the output stage. Moreover, a CCR power supply combined with grid bias allows us to set the quiescent current of the tube and we can now vary the grid voltage to obtain the best sonic results without changing the quiescent current. The plate voltage will float to the necessary value for the tube to conduct the set current at the set grid voltage.

I took advantage of the CCR / grid bias arrangement to make changing the operating conditions between a 50 or a 300B a simple matter of turning a switch. I adjusted the grid bias voltage values for the 50 and 300B such that the distortion residual (monitored on a ‘scope using a HP339a analyser) was cleanly 2ND harmonic across the full dynamic range of the amplifier. The filament current draws of the 50 and the 300B are very nearly equal so I used current regulation for the filament also, thereby avoiding the need to change the filament supply setting when changing tube types. I intend to write this design up fully in a future article so I won’t say more other than it sounds excellent, more open and able to resolve far more deeply than my previous designs and without false modesty, than takes something special!


Figure 7 shows two push-pull topologies that use CCR technology. Referring to figure 7A, the CCR is located in the plate circuit so there is no cathode coupling; the circuit exhibits current balance but not current and voltage balance. Taking an extreme case, if we apply the input to one grid only, the result will be gross distortion. We can do much better if we place the CCR in the cathode circuit, see figure 7B; the circuit is now a long tailed pair that is completely self balancing, fully compensating for any imbalance in the input signals.


The first approach I came up with is the “Current Balanced Push-Pull” amplifier, published in the May 2006 edition of Audioxpress. When I created the current balanced amplifier, I was primarily exploring the potential of inexpensive toroidal power transformers as output transformers. The result is one of the best sounding amplifiers I have built or have heard. This implementation originally used a simple (single) mosfet type CCR located in the cathode of each output triode to assure near perfect DC current balance so as to avoid DC saturation of the toroidal output transformer; the cathodes are coupled together with a capacitor to obtain perfect signal balance in the output tubes and thus true Class A; I dubbed this arrangement a “Power Long Tailed Pair”. Because the push-pull signal path loop impedance is higher than the cathode resistance of the triodes, (refer to the chain-dashed loop on figure7) the capacitor does not need to be in the range >100 µF, in fact in this case it is a 10 µF film type. The effectiveness of the CCR technology is revealed by the fact the output and drive stages share a common supply; I and my friend, bassist John Dahlman found it difficult to distinguish if a PSU bypass capacitor was present or not, suggesting that the true Class A push-pull behaviour made possible using CCR technology frees us from PSU bypass capacitor quality concerns. I recently updated this design, replacing the simple mosfet CCRs with cascode CCRs, see figure 8.


Many years ago, John Camille developed a single-ended 845 amplifier. He courageously shunt regulated the B+ supply using another 845 controlled by the signal via an op. amp. I am clear that well implemented shunt voltage regulation is superior to series voltage regulation; for one thing, the shunt may be fed using a CCR thereby isolating the shunt signal loop from the PSU. Another characteristic of the shunt regulator is that it can both feed and absorb current, perfect for output stages. However John’s design and until recently, other implementations I have seen are asymmetrical; the shunt circuit does not complement the signal circuit.

I am beginning to wonder if true Class A push-pull may actually be superior to single ended, gasp! I suspect that many so-called Class A push-pull amplifiers actually cross into Class AB during more of the program material then is generally recognised with the result that the not only is the PSU modulated audibly but also, the output resistance of the amplifier is continuously switching. In past days, 20 dB of negative feedback would most likely have swamped this issue (and the music) but with ‘modern’ zero feedback triode designs; I suggest that this issue might horribly compromise the sonic results. Whether this is valid or not, true Class A push-pull opens up the possibility of a self-shunt regulating amplifier.

The point I am making about a true Class A push-pull amplifier is that given that it is fed via a CCR, (preferably in the cathodes to assure near perfect signal current and voltage balance) then it is self-shunt regulating, refer back to figure 7B. And so for me, the next level is to take what I have started with the 6AS7G output and drive stages and apply this self shunt-regulation, true Class A push-pull technology to all the stages. In fact, Kevin Carter of K&K Audio is doing this and his GM70 PP design is the best sounding amplifier I have yet heard. Furthermore, Kevin has nice PCBs for building CCRs available.


After a year or so, my 6AS7G PP amplifier failed due to overheating of the DN2540 mosfets used in the CCRs: The current levels went far out of specification resulting in the amp going from extremely musical to horrible! Part of the problem was due to inadequate ventilation. However, I had been considering replacing the simple CCRs with cascode units for a while and this was the opportunity! Referring back to figure 2, you will see that in this topology, the VDS of the critical control mosfet M1 is limited by the VGS of the cascode M2: Clearly this topology has the further advantage of causing very low heat dissipation of the critical control mosfet, M1. If the voltage drop is large, I take some of the thermal load off the CCR by putting an appropriate value of power resistor in series with it. In this case, I spread the 6AS7G cathode voltage 50/50 between the CCRs and resistors. I mount the two mosfets back to back on a single heatsink, however I now use larger heatsinks than I think are necessary and ensure that they are in free air. Moreover, given that the mosfets are back-to-back I use two mica insulators and no thermal grease on M1 to reduce the heat soak from M2 and a single mica washer with thermal grease on M2. So far, this is working well, the dc current in the output tube push-pull sections has remained stable and the current balance between the push-pull sections remains better than 0.5 mA.


Other possible applications of the depletion mosfet two-terminal CCR technology are:

1/ Norton dc signal level shifter;

Sometimes it is desirable to shift the DC level of the signal, for example when DC coupling. Figure 9 shows a Norton level shifter that simply superimposes a clean DC current on the signal; the DC current flows through a resistor interposed between the plate of the first stage and the grid of the second stage causing a DC potential difference that can be arranged to match the potential difference between the plate and the grid. Since the CCR has an impedance of 1.5 MΩ or higher, it adds little load to the previous stage. The operating conditions of the circuit need to be extremely stable (as with all DC coupled circuits) so PSU voltage regulation is a good idea. Also, the value of the interposed resistor R will interact with the Miller capacitance of the second stage. All in all, I think this device is mostly of curiosity value, especially given that interstage transformer coupling sounds so excellent. Still, one of you may find a use for it!

2/ Precision variable ultra low-noise DC voltage reference;

I still use voltage regulation in some areas, for example, my SE 50-300B design uses mosfet series regulation ahead of the CCRs. I wanted a simple way to adjust the output voltages so zener diodes were not the best choice for the gate voltage reference. Since the DN2540 cascode CCR is so extremely stable, I simply built an adjustable one (a pot for the SOT resistor in figure 2) feeding current through a thermally stable metal film resistor (having an adequate voltage rating) to obtain a precise, ultra clean reference voltage that does not drift around as things get warm, (unlike a zener).

3/ Tuning the load of a pentode or cascode to obtain the best sonic results and precisely define the driving impedance for a Lipshitz (or other) passive RIAA equalisation network;

This is a nice one and I almost hesitate to give it away, see figure 10.

I have found that a hybrid, jfet–triode cascode works and sounds extremely well for the front-end of a phono stage.

So far, I am not alone with this experience. However, the next issue is providing well-defined source impedance for the following Lipshitz (or other) passive RIAA equalisation. The trick is to use a CCR to feed the current to the plate of the cascode; it only needs 15 V to work. Then put a resistor in parallel with the CCR to set the load. The resistor value may be adjusted to provide the best harmonic characteristics from the stage without changing the plate voltage or B+ voltage required unless the resistor is so small that it starts to conduct appreciable DC current; in that case, a CCR is redundant. You can use a pot for the resistor while experimenting; when the best value has been found, replace it with a fixed resistor. Now, the plate load resistor value controls the cascode output impedance; the cascode plate resistance in my designs at least, will be around 3 MΩ so any variation in the cascode RP with age will have negligible impact on the accuracy of the equalisation since 3 MΩ is likely to be significantly greater than the value of the plate load resistor.

4/ Shunt regulator current feed;

If voltage regulation is desired – and it can be a superior way to implement ultra high-resolution, small-signal stages such as phono-stages – then arguably the best way to do it is shunt regulation. Figure 11 shows the topology of the stage specific shunt regulators I created for my WE 416C MKII phono-stage.

The shunt path is shown by the chain-dashed loop and again, the CCR isolates the shunt path completely from the PSU with sonic benefit. Since this is a no-compromise design, I used a negative supply to permit the shunt triode to be directly across the phono-stage supply with nothing in the cathode circuit. Note; in this case, the shunt regulator is in the signal path; a corollary to this is that an individual shunt regulator must be used for each stage to realise the resolution benefits of shunt regulation. I have found that the effect of bypass capacitors in my WE 416 phono-stage are inaudible to me (I actually can switch them in and out of circuit to compare), so I am listening to the regulators; this is the benefit of ensuring that a shunt regulator design is a match for the amplifier stage design, not a generic design. To reiterate, I suspect that the best shunt-regulated audio circuit is a long tailed pair having a CCR as the tail.


My intention in writing this is to illustrate the potential for the thoughtful application of solid-state devices to more fully realise the sonic beauty of vacuum tubes than was possible without such devices. It is noteworthy that Western Electric had envisaged (maybe originated) the use of a CCR in the PSU current feed to a parallel feed, single ended output stage and now, we can easily experiment with this concept.

I hope you find the ideas expressed in this article stimulating and applicable. I am a huge fan of experimentation; that is where the fun is so I encourage you to play!

Thank you for your attention.

Richard Sears, April 2010

1/ Walt Jung; Letter to Audioxpress, April 2009

2/ Loop Distortion; Lynn Olson,

3/ Ultrapath, Parallel Feed and Western Electric; Lynn Olson, Vacuum Tube Valley, Issue 16

Source for CCR kits; K&K Audio (Kevin Carter),

Source for Supertex mosfets; Mouser Electronics,

Source for IXYS mosfets; Arrow Electronics, http://www


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  1. Superb! usually We by no means study whole content articles however the way a person authored this post is simply amazing this also kept my personal consciousness in analyzing and that i enjoyed it.

  2. Vyacheslav permalink

    Richard, this is a very big job, you’ve done a lot of research, many measurements, analysis! This is very valuable information! accessible, clearly written. Сurrently – this is the most complete information about the current source for the amplifiers in the Internet!
    Thank you. We are waiting for your new studies and by measurement and new posts on these themes!
    Ukraine, Dnepropetrovsk.

    • Dear Vyacheslav.
      Thank you for your comments. You are actually the only person that I know of, who gets what I have done. It was intended as a contribution to the audio community and it did not “land”.
      As for further work, at this point I doubt if there will be any. My thoughts and interests have moved on. I hope you can find some value in application of the thoughts I presented and discussed.
      Richard Sears

  3. Vyacheslav permalink

    you have new projects on the amplifier?
    I saw your amp to 50-300B SE ( – an excellent amp. the idea of ​​”isolate the audio circuits from the power supply” – is very good.
    I have not found Similar designs on the Internet and in literature! 🙂
    I want to build for yourself like that! 🙂
    Sorry, that I write to you here, on http://www.triodeguy.tsom can not leave comments.
    Good luck in your endeavors!
    Ukraine, Dnepropetrovsk.

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